Video transmission system

ABSTRACT

A subscriber cable television system uses predominantly digital signal processing techniques and has extremely high security and an increased capacity for transmitting program and customer data to individual decoder units. For ease of data handling, two-channel audio, video, and high capacity program and customer data are multiplexed for transmission on the composite video signal. The decoder unit employs a system timing circuit which precisely synchronizes the sample times on the received composite video signal to the chroma burst, regardless of whether the video information is for a color or black-and-white program. An improved time-warp and segment scrambling method is disclosed along with means for suppressing the undesirable effects of discontinuities in the scrambled video signal. The digital audio is transmitted as scrambled most significant bits in low resolution samples and unscrambled least significant bits in a high resolution remainder sample. The system timing circuit has a horizontal sync detector accommodating variable line length such as is provided by some video recording apparatus. The clock to the horizontal counter is selectively phase-reversed in response to early or late horizontal sync so that the timing resolution is twice the clock period. An improved self-adjusting threshold detector and other means are disclosed for detecting a &#34;20 IEEE&#34; suppressed horizontal sync so that the full range of video modulation may be used more effectively. Circuitry is also disclosed for transmitting the customer and program information in a multi-level correlative signalling format in order to more effectively use the band width of the entire television channel.

This is a of co-pending application Ser. No. 564,405 filed on Dec. 22,1983 U.S. Pat. No. 4,682,360.

BACKGROUND OF THE INVENTION

This invention relates generally to secure communication systems and,more particularly, to cable television systems wherein designatedsubscribers are enabled to receive particular program material.

In any such subscriber television system, means are required forscrambling the audio and video information, and means are also requiredfor transmitting program and subscriber information to designatesubscribers permitted to view particular programs. Although the priorart discloses a wide variety of methods, commercially accepted systemstypically perform these functions by independent analog scrambling ofthe audio and video information, and by multiplexing digital customerand program data with either the audio or video signals. Populartechniques include, for example, "sine-wave scrambling" of the video tosuppress horizontal synchronization and modulating the audio informationon a supersonic subcarrier. More advanced commercial systems employ apseudo random code for scrambling the video information, for example, bypolarity inversion of the video signal on a frame-by-frame orline-by-line basis. These techniques have required increased complexityand cost, but the additional security is needed to frustrate pirates whohave gained considerable skill and experience in circumventing securitymeasures.

SUMMARY OF THE INVENTION

A principal object of the invention is to provide a subscriber cabletelevision system having an inexpensive decoder yet also havingextremely high security and an increased capacity for transmittingprogram and customer data to the decoder units.

Another object is to provide a video transmission system whereintwo-channel audio, scrambled video, and high capacity program andcustomer data are all multiplexed onto a composite video signal.

An additional object is to provide an improved method of transmittingdigitally-scrambled audio.

Still another object is to provide an improved system timing circuit inthe decoder unit for defining precisely synchronized sample times on thereceived composite video signal.

Yet another object to provide a horizontal sync detector circuit capableof accommodating variable horizontal rates, such as are generated bysome video recording apparatus.

And another object is to provide a method of synchronizing a counter toa reset pulse to obtain a timing resolution exceeding the clockingperiod of the counter.

Moreover, it is an object to provide means for using the full range ofvideo modulation for tranmsitting the video portion of the compositevideo signal.

BRIEF DESCRIPTION OF THE DRAWINGS

Other objects and advantages of the invention will become apparent uponreading the following detailed description and upon reference to thedrawings, in which:

FIG. 1 is a block diagram of the encoder or scrambler portion of thevideo transmission system according to the invention;

FIG. 2 is a block diagram of the decoder or descrambler portion of thevideo transmission system;

FIG. 3 is a pictorial diagram of one horizontal line of the video signalencoded by the encoder of FIG. 1;

FIG. 4 is a pictorial diagram showing the contents of the scrambledvideo or customer data portion of the horizontal lines by line number inthe television frame;

FIG. 5 is a pictorial diagram of the horizontal synchronization andchroma burst portion of the beginning of each encoded horizontal line;

FIG. 6 is a pictorial diagram showing the arrangement of thehousekeeping data packed into the first horizontal line of thetelevision frame;

FIG. 7 is a pictorial diagram showing the record number and parity checknumber in each line of customer data in time-slot format;

FIG. 8 is a block diagram of the decoder or receiver of FIG. 2, showingin detail the system timing portion which is common to both the encoderand decoder of the video transmission system;

FIG. 9 is a pictorial diagram showing the method of scrambling orencoding the video portion of the horizontal lines to preventunauthorized reception;

FIG. 10 is a simplified block diagram of a one bit portion of the randomnumber generator used in the encoder and the decoder for preventingunauthorized persons from determining the encoding key merely bydetailed analysis of the encoded signal;

FIG. 11 is a pictorial diagram showing the undesirable effects ofdiscontinuities introduced by the encoding scheme and one method forreducing the distortion from the decoded video signal;

FIGS. 11A, 11B, and 11C are pictorial diagrams showing the preferredmethod for virtually eliminating the distortion caused by thediscontinuities introduced during scrambling;

FIG. 12 is a pictorial diagram showing the memory contention problemoccurring when the total number of video samples in a video line isincreased and decreased during video encoding and decoding;

FIG. 13 is a schematic diagram of a video scrambler/descrambler having a256 byte memory in order to avoid the memory contention problem;

FIG. 14 is a pictorial diagram showing the sampling and encoding oftwo-channel audio;

FIG. 15 is a block diagram of the audio scrambling circuits for thevideo transmission system;

FIG. 16 is a block diagram of the audio descrambling circuits for thevideo transmission system;

FIG. 17 is a schematic diagram of a chroma burst phase-locked loop;

FIG. 18 is a schematic circuit diagram of a horizontal sync detectoraccommodating varying line length and tolerating missing horizontal syncpulses;

FIG. 19 is a timing diagram depicting the window gate signal whichqualifies horizontal sync pulses;

FIG. 20 is a schematic circuit diagram of a horizontal sync tipthreshold detector;

FIG. 21 is a schematic circuit diagram of a self-adjusting thresholddetector having a single operational amplifier functioning as a peakdetector and as a threshold detector;

FIG. 22 is a pictorial diagram of the input and output signals for thethreshold detector of FIG. 21;

FIG. 23 is a pictorial diagram of the operation of the thresholddetector of FIG. 21 when processing a video signal having a "20 IEEE"suppressed horizontal sync;

FIG. 24 is a schematic circuit diagram of a circuit for detecting "20IEEE" suppressed horizontal sync;

FIG. 25 is a schematic circuit diagram of a correlative encoder forefficiently transmitting the digital customer and program data withinthe bandwidth of the television channel;

FIG. 26 is a schematic circuit diagram of the decoder for the encoder ofFIG. 25, including error detection circuitry;

FIG. 27 is a timing diagram for a clock rephase circuit for permittingthe horizontal counters to have increased timing resolution;

FIG. 28 is a schematic diagram of a clock rephase circuit generating thesignals shown in FIG. 27; and

FIG. 29 is a schematic diagram of a fast acting clamp circuit forestablishing a ground reference level for the received video signal.

While the invention is susceptable of various modifications andalternative constructions, a certain preferred embodiment has been shownin the drawings and will be described below in detail. It should beunderstood, however, that there is no intention to limit the inventionto the specific form described but, on the contrary, the intention is tocover all modifications, alternative constructions and equivalentsfalling with the scope of the appended claims.

DESCRIPTION OF THE PREFERRED EMBODIMENT

Turning now to the drawings, there is shown in FIG. 1 a block diagram ofa scrambler or encoder generally designated 20 for one embodiment of thevideo transmission system of the present invention. The scrambler 20receives station video on an input 21, two-channel station audio signalson an input 21' and customer and program data on a digital input 21".The scrambler 20 is most generally described as a synchronous logiccircuit, with a predetermined set of operations occurring periodicallyat predefined times with respect to the vertical and horizontalsynchronization signals present in the station video on the input 21. Inorder to synchronize the scrambling functions with the horizontal andvertical synchronization signals, sync detectors 22 isolate thehorizontal and vertical synchronization signals from the station video,and a chroma burst phase-locked loop 23 generates a system clockprecisely phase-locked to the chroma burst signal in the station videoso that the operations of the scrambler 20 may be even more preciselydefined as a function of time with respect to the horizontal andvertical synchronization signals. The precise beginning of a horizontalline, for example, is indicated by the first transition in the systemclock following a horizontal sync edge, thereby rejecting noise orjitter in the position of the horizontal sync edge. The detectedhorizontal and vertical sync signals are used to reset horizontal andvertical state counters 24 having decoders for generating gating signalswhich enable the scrambling or encoding operations to be performed atprecise predetermined times with respect to the horizontal and verticalsynchronization signals. A chroma burst gate, for example, is fed backto the chroma burst phase-locked loop 23 to specify the time at whichthe chroma burst signal is present in the station video on the input 21.

The scrambling of the video signal is performed on a time-sampled basis,and the video samples are processed in digital form. The station videoon the input 21 is sampled at three times the frequency of the chromaburst by an eight bit analog-to-digital converter 25. Since the chromaburst frequency represents the frequency of the suppressed carrier forphase modulation of the chroma signals, a sampling rate of three timesthe chroma burst frequency is about the lowest sampling rate that may beused. For the standard 3.58-MHz chroma burst frequency, the sampling ofthe eight bit analog-to-digital converter 25 occurs at a 10.7 MHz ratein synchronism with the system clock from the chroma burst phase-lockedloop 23. Because of this high sampling frequency, the analog-to-digitalconverter 25 is a parallel mode or "flash" converter. The digital videooutput of the analog-to-digital converter 25 is quantized to 256 levelsspecified by the eight bits from the analog-to-digital converter; inpractice it is found that the eight bits are sufficient for generating adigital video signal which may be processed and converted back to analogform without significant visual perception of the quantization error.Since the sampling rate is a multiple of the chroma burst frequency, thequantization error components coincide with the existing components inthe television signal, and the intermodulation products "zero beat" tobecome unobtrusive.

Due to the delay associated with video scrambling, sync and chroma burstregenerating circuits 26 are used to generate, in digital form, adelayed version of these signals. This delayed version is preciselysynchronized to the system clock, and is in effect pre-programmed in amemory addressed during the beginning of each encoded video line.

A video scrambler 27 receives the digital data encoding the beginning ofeach horizontal line from the sync and chroma regenerating circuits 26.Then the digital video is encoded in the video scrambler 27 by randomlyscrambling the time positions of the digital video samples in responseto a predetermined random number. The random number is specified by afree-running random number generator 28 containing predetermined andsecret key logic. A seed number generated by the random number generator28 is periodically transmitted from the encoder 20 to the decoder (40 inFIG. 2) to maintain synchronization so that a copy of the random numbermay be generated by a similar random number generator in the decoder.Because the key logic is secret, the random number cannot be generatedmerely from the seed which could possibly be intercepted as it istransmitted from the encoder to the decoder.

The seed numbers, as well as digital data for transmitting programidentification and customer data, are time multiplexed with thescrambled video information and transmitted from the encoder to thedecoder. The digital data, for example, are received from the customerand program data input 21" and loaded into storage registers in thedigital data formatting circuits 29. These buffered or stored data arethen transmitted at predetermined times in response to gating signalsfrom the state decoders 24.

The two audio channels are also digitized, scrambled, and timemultiplexed with the scrambled video and the customer and program data.Each channel of the two-channel station audio 21' is fed to a separateone of two audio digitizers 31, each generating binary audio data at arate of 36 bits per horizontal video line, or 565,200 bits per second.The binary audio data for each audio channel for each horizontal lineare scrambled by an audio scrambler 32 in response to the random numberprovided by the random number generator 28.

In accordance with one feature of the present invention, the digitalaudio and the customer and program data are formatted into two-bitbinary numbers which are converted to analog samples and then timemultiplexed with the scrambled video for transmission. The 72 binarybits of audio for each horizontal line are formatted into 12 two-bitbinary numbers and six eight-bit binary numbers containing lessersignificant digital audio bits. The binary numbers are routed by anoutput multiplexer to an eight-bit digital-to-analog converter 33 sothat the analog samples are placed before the video portion of eachhorizontal line in the encoded composite video. Similarly, the digitizedcustomer and program data are also transmitted as analog samples.

Due to the limited band width of the transmission channel, the scrambleddigital video cannot be transmitted through the channel in digital form.Therefore, the eight-bit digital-to-analog converter 33 is required togenerate an analog video signal from the time scrambled digital videosamples.

In accordance with another feature of the present invention, a chromaburst signal is always transmitted regardless of whether the stationvideo is for a color or black and white program. The chroma burst signalis used at the decoder for generating a system clock synchronized to thesystem clock in the encoder. Thus, the sample times for the audiosamples and the customer and program data may be determined at thedecoder from the system clock phase-locked to the received chroma burst.

A block diagram of the descrambler or decoder 40 is shown in FIG. 2. Theencoded composite video is received on an input 41, and sync detectors42 strip off the horizontal and vertical synchronization information.The chroma burst phase-locked loop 43 regenerates a system clock and thesystem clock is fed to state counters 44 being reset by the horizontaland vertical synchronization signals. As in the encoder 20, the statecounters have decoders for generating gating signals such as a chromaburst gate which enables the chroma burst phase-locked loop 43 to sensethe received encoded composite video. The received encoded compositevideo is also fed to an eight-bit analog-to-digital flash converter 45for generating a digital video signal.

In accordance with another feature of the present invention, thereference for the eight-bit analog-to-digital converter 45 in thedecoder 40 is not supplied by a peak detector, but rather the referenceis provided by a reference detector 46 gated by reference gates from thestate counters and decoders 44. The eight-bit digital-to-analogconverter 36 in the encoder 20, for example, outputs a maximum or "max"reference signal in response to a max reference gate from the statecounters and decoders 24 so that a precise full scale signal istransmitted from the encoder 20 to the decoder 40. The state countersand decoders 44 in the decoder similarly generate a max reference gatefed to a max reference detector 46 which samples and holds the receivedmax reference signal, for use as a reference to the eight-bitanalog-to-digital converter 45. The state counters and decoders 44 alsogenerate a ground reference gate for setting the horizontal sync tipsprecisely at signal ground so that the range of the eight-bitanalog-to-digital converter 45 is fully determined.

The digital video is fed to a video descrambler 47 and digital dataprocessing circuits 49. The digital data processing circuits strip theseed number from the digital video and pass it to a random numbergenerator 48. The seed number synchronizes the random number generator48 in the decoder 40 to the random number generator 28 in the encoder20. The random number regenerated in the decoder 40 is passed to thevideo descrambler 47.

A customer identification PROM 50 is programmed with a unique customernumber for each decoder. The customer identification number specifies atime slot in the digital video signal when specific customer informationis being transmitted and received. Some of this information is availableon an output bus 51 so that the video transmission system may providedigital data transmission to support future products such as teletextand electronic mail.

An audio descrambler 52 also receives the digital video signal and therandom number from the random number generator 48. The two-channel audiosignals are regenerated by two separate digital-to-analog converters 53and are fed via output lines 54 to the customer's stero hi-fi. Twochannels rather than just one are desired to provide for transmission ofstereo or bi-lingual programming. The descrambled video, on the otherhand, is converted to analog form by an eight-bit digital-to-analogconverter 56, and a combiner 57 mixes the descrambled video with signalsfrom an NTSC sync, blanking, and chroma burst generating circuit 58 sothat an NTSC composite video signal results. The NTSC video is fed tothe customer's television set. It should also be noted that the NTSCsync, blanking, and chroma burst circuits 58 receive a color killersignal from the digital data processing circuits 49 in order that thechroma burst is removed from the NTSC video signal when ablack-and-white program is being received. Conventional television setshave color killer circuits to turn off the chroma amplifiers when ablack-and-white signal is being received, and these color killercircuits are responsive to the presence or absence of the chroma burstsignal. Thus, the video transmission system according to the presentinvention transmits a digital signal from the encoder to the decoder sothat the NTSC video signal fed to the customer's television set may havethe chroma burst deleted to improve the reception of black-and-whiteprograms.

A pictorial diagram of the transmitted encoded composite video signal,generally designated 60, is shown in FIG. 3. The signal is drawn so that"white" is at the top or maximum of the video signal 60 and "black" isat the bottom of the video signal. The conventional horizontal sync tip(shown in dashed representation), in fact, is more negative than thevideo signal for the darkest picture element. Thus, the horizontal synctip is typically stripped from the rest of the video signal by athreshold detector set at approximately the "black" level. In theapplicant's preferred embodiment, however, the conventional horizontalsync tip is suppressed as shown, and the circuits shown in FIGS. 22-25and described below are used to in effect regenerate the suppressedhorizontal sync. Following the suppressed horizontal sync tip is thereference pulse at the maximum of the video signal. Following the maxreference pulse is the chroma burst of approximately 3.58 MHz. Thechroma burst is 100% peak-to-peak about a 50% video level in order toreduce phase-lock noise susceptibility. Following the chroma burst aretwo digital samples containing vertical synchronization information.Each digital sample is comprised of two sample clock cycles since thefirst sample clock cycle is used for slewing from one digital sample tothe next. The digital samples have a maximum value, for example, toindicate that the next horizontal line is the first line in the videoframe, and have a minimum value otherwise. Following these digitalsamples are 18 digital samples of digital audio comprising a total of 36sample clock cycles. The horizontal line is completed by 590 sampleclock cycles of scrambled video or customer data.

A pictorial diagram of the information content for one frame 65 of theencoded composite video signal is shown in FIG. 4. The horizontal linenumbered 0, immediately following a vertical sync pulse, containsdigital housekeeping data. Horizontal lines 1-8 contain customer data ina time-slot format. The data for a particular customer, for example, arelocated at a particular position in a particular one of the horizontallines 1-8. Lines 9-260 contain the scrambled video for the even field ofthe frame 65. Horizontal lines 261-270 continue with customer data intime slot format. Finally, lines 271-524 contain the odd field of thescrambled video in the frame 65. It should be noted that the verticalsync and digital audio have been placed in the horizontal blanking timeof the 63.5 microsecond horizontal line. Similarly, housekeeping andcustomer data in time slot format have been inserted in the conventionalvertical sync and blanking time of the video frame. These sync portionsand conventionally unused blanking portions of the video signal can bepacked with data since synchronization for both color andblack-and-white programs is based on the use of the chroma burst, andthe blanking signals are regenerated by the decoder using the NTSC sync,blanking, and chroma burst generating circuits 58 (FIG. 2).

The acquisition of horizontal sync is illustrated by the timing diagramin FIG. 5. Each of the chroma burst phase-locked loops 23, 43 comprisesa crystal oscillator having a fundamental frequency of six times thechroma burst frequency, or approximately 21.4 MHz. This 21.4 MHzfrequency is used as a system clock to define the times for operation ofthe various logic functions in the encoder or decoder. The system clockfrequency is divided by two in order to generate a 10.7 MHz sample clock70 for digitally processing the video signal. The sample clock 70 is theprimary phase reference for all decoding operations.

The first falling transition of the sample clock after the leading edge60a of the horizontal sync tip, for example, defines the beginning of ahorizontal reset pulse 71a having a width of one sample clock. Thehorizontal reset pulse is used as a synchronous reset input to ahorizontal counter clocked by the sample clock to define a horizontalposition or count along each horizontal line. In other words, the timezero reference for each line is a predetermined phase point on thedecoder phase reference immediately after the horizontal sync edge. Achroma burst gate 72, for example, enables the chroma burst phase-lockedloops 23, 43 during horizontal counter states 20-46. A max referencegate 73a is enabled for horizontal counter states 11-13 in order tosample the top of the max reference pulse, and a ground reference gateis enabled for horizontal counter states 2-5 to set the horizontal synctip at signal ground. In a similar fashion, gating signals for thevertical sync, the digital audio, and the scrambled video or customerand program data are generated from the horizontal counter states. Thusthe time positions of the analog samples in the composite video encodingthe audio information and customer and programming data are synchronizedto the decoder phase reference at predetermined time intervals from thetime zero reference.

In practice, for example, a programmable logic array (PLA) has outputsthat are programmed to be active upon various horizontal counter states.A gating signal which is logically high or low for a number of cycles isconveniently generated by a JK flip-flop with its J input being activeupon the horizontal counter state preceding each low-to-high transitionof the gating signal, and its K input active upon the horizontal counterstate preceding each high-to-low transition of the gating signal. Itshould also be noted that in general the horizontal and verticalcounters are preset to initial states different than zero. In "state 0"of FIG. 5, for example, the horizontal counter should have a largeinitial value so that the counter "rolls over" to zero when thescrambled video portion of the horizontal line is reached. Then theoutput of the horizontal counter can be used to directly address videomemory in the video descrambler 47 (FIG. 8). It is also desirable topreset the horizontal counter more than once per horizontal line. Thetrailing edge 60b of the horizontal sync tip, for example, is a moreprecise and stable time zero reference for the horizontal line since thetrailing edge is a transition from the most negative to the mostpositive portion of the composite video signal. Thus it is desirable togenerate a horizontal preset pulse 71b synchronized to the sample clockimmediately after the trailing edge 60b. The horizontal preset pulse 71bthen presets the horizontal counter to state 9.

A pictorial diagram of the customer and program data in the housekeepingline 74 is shown in FIG. 6. The line contains, for example, a systemidentifier, program information, the random number generator seed, andmiscellaneous or reserved data slots. The system identifier is, forexample, a 24-bit number unique to the system, allowing over 16 milliondistinct systems. Reception of the assigned system identifier may beused to enable the decoder so that units stolen from one system cannotbe used secretely in another system. In other words, the systemidentifier is a "global prefix" to the customer I.D. numbers.

The program information identifies the video information or programmingbeing viewed. A "parental guidance" rating is the most significant partof the program information. It is desirable to have a number of ratinglevels, for example sixteen specified by a four-bit number. To enhanceparental control over the TV reception, a threshold level could betransmitted from the control station to the individual decoders tospecify the highest rating of programs to be decoded.

Another important part of the housekeeping data is the seed for therandom number generator 48. To maintain synchronization of the randomnumber generator 48 (FIG. 2), the seed is loaded into the random numbergenerator for each video frame.

Preferably a few locations on the housekeeping line are reserved formiscellaneous functions. For enhanced security, these miscellaneousfunctions could include changing the key logic or the format for theloading of the random number seed. Another desirable miscellaneousfunction is to change the customer data from a time-slot format to astream format wherein the customer data in time slot format is used toenable a group of customers and then the customer data is transmitted instream format to all of the enabled customers. The stream format modecould be used, for example, for transmitting electronic mail to theselected group of customers.

Shown in FIG. 7 is a line 75 of customer data in time-slot format. Itshould be noted that all of the customers cannot be individuallyaccessed by transmitting customer data in time-slot format during just asingle line of customer data. Thus, the group of customers selected forreceiving customer data in time-slot format must be identified by arecord number preferably appended to the beginning of a correspondingline 75 of customer data. A 16-bit record number, for example, specifiesone of 65,536 different groups of customers having time slotscorresponding to the same record number. A maximum cycle time of abouttwo seconds is required for individually addressing over one-halfmillion subscribers. The record number associated with a particularcustomer is, for example, the prefix or most significant bits of thecustomer I.D. number in the customer I.D. PROM 50 (FIG. 2). Certainrecord numbers could be reserved for addressing groups of customers. Arecord number of 0, for example, could indicate that every customer isto receive the customer data in time-slot format for the current line 75of customer data. In this case, when the record number is 0, thecustomer information detector 92 does not require a match of the recordnumber with the most significant bit portion of the customer I.D. numberin the customer I.D. PROM 50. Then the customers would be groupedaccording to their common time-slot position in the frame. This methodof grouping would permit any selected group to be accessed within lessthan half the framing period, or within less than approximately 17milliseconds. The customers are accessed on an individual basis, forexample, to completely disable the decoders for customers who have notpaid their bills, or to enable the accessing of special programmingmaterial for individual customers. The parental guidance threshold levelfor each customer, for example, is periodically transmitted as customerdata in the corresponding time-slot positions. Customers could beaddressed in a group format, for example, according to geographiclocation for the transmission of local news.

An important feature of the customer data line 75 is a parity checknumber at the end of the line. Since a parity check is used, the matchof the beginning of the video portion of any line having the particularrecord number can start the detection of customer data at the time slotposition, since the parity check may be used to exclude or ignorecustomer data received when the record number is falsely generated bythe scrambled video. Thus customer data may be transmitted at high speedby temporarily interrupting the transmission of scrambled video andtransmitting customer data in time-slot format in every video line,including vertical sync and blanking. The decoding of customer data isalso simplified since the operation of the decoding process isindependent of the line number of the line 75 of customer data.

A more detailed block diagram of the descrambler or decoder is shown inFIG. 8. A horizontal sync tip window detector 81 strips the horizontalsync tips from the received encoded composite video on the input 41, andthereby isolates the leading and trailing sync edges 60a, 60b. Ahorizontal sync detector 82 generates the horizontal preset pulse 71b insynchronism with the 10.7 MHz sample clock immediately following thetrailing sync edge 60b. An additional feature to be described completelybelow is that the horizontal sync detector 82 will generate a presetpulse even if a horizontal sync tip is not received. The horizontal syncdetector 82 will estimate when the horizontal preset pulse should occurbased on the previous horizontal line. The preset pulse from thehorizontal sync detector 82 presets a horizontal counter 83 clocked bythe sample clock to generate a horizontal sample position or count. Thestate decoder PLAs 84 generate a chroma burst gate, a vertical syncgate, a max reference gate, and a ground reference gate as a function ofthe horizontal sample position, as well as other gating signals. Thevertical sync gate is fed to a vertical sync detector 85 to generate avertical reset pulse V fed to a vertical counter 86. The verticalcounter is a synchronous counter clocked by the sample clock but enabledonly once for each horizontal line by a horizontal pulse H generated bythe state decoder PLAs 84. The horizontal line number generated by thevertical counter 86 is also fed to the state decoder PLAs to generategating signals which enable the processing of data formatted accordingto horizontal line number, as shown in FIG. 4.

The max reference gate is fed to the max reference detector 46 shown asincluding a sample and hold circuit having an analog transmission gate87a, a holding capacitor 88, and a buffer 89. The buffer 89 feeds thereference of the eight-bit analog-to-digital converter 45. The groundreference, on the other hand, is obtained by a peak detecting orsampling circuit. The ground reference is subtracted from the receivedcomposite video to restore the DC level of the video signal. A preferredcircuit is described below in conjunction with FIG. 29. Alternatively, atransmission gate 87b sets the ground reference.

The two most significant bits of the digital video output of theanalog-to-digital converter 45 are fed back to the vertical syncdetector 85. All eight bits are fed to the video scrambler 47 and theaudio descrambler 52. Only the two most significant bits are fed to therandom number seed detector 91, and a customer information detector 92.The customer information detector 92 is responsive to the housekeepinggates and is enabled at the particular customer data time-slotcorresponding to the customer identification number stored in thecustomer ID PROM 50. A digital comparator 93 generates the customer gatesignal when the least significant bits in the customer ID PROM 50 equalthe customer gating time specified by the horizontal counter 83 and thevertical counter 86. The customer ID PROM 50, in other words, includesthe particular horizontal count or counts corresponding to the assignedtime slot in the customer data time-slot positions shown in FIG. 4, aswell as the particular record number assigned to the customer.

The customer information detector 92 includes the gated latches whichreceive the customer and program data. Separate latches, for example,store the record number received from the beginning of the video portionof every line, the 4-bit parental guidance threshold for the particularcustomer, and the 4-bit parental guidance rating for the particularprogram. The latch for the parental guidance threshold is enabled by thecoincidence of the customer gate from the digital comparator 93,specifying the customer's time-slot position in the frame, and thematching of the output of the record number latch and the mostsignificant portion (MSB) of the customer identifier stored in thecustomer ID PROM 50. The customer information detector 92 generates anenable signal fed to the random number generator 48 to permit programreception. The enable signal is active unless the numerical value of theoutput of the parental guidance rating latch exceeds the output of theparental guidance threshold latch, as determined by a 4-bit digitalmagnitude comparator. Note that the parental guidance threshold andrating may be used to effectively turn off non-paying customers, forexample, by transmitting a threshold of zero to non-paying customers andassigning the rating of zero just to commercials and channelidentification or emergency messages. The highest rating of fifteen, ofcourse, is reserved for the most sensitive and restricted adult programmaterial.

Also shown in FIG. 8 is a monochrome detector 94 receiving the two mostsignificant bits of the digital video for generating the color killersignal. It should be noted that the detector circuits in general aresynchronous latches enabled at the respective gating times. A typical ICpart number is 74173. The corresponding logic in the encoder of FIG. 1is even more simple since the vertical pulse, monochrome signal, andother data are merely multiplexed onto the output of the digital dataformatting circuit 29 at the corresponding gating times.

Since it is especially important that a vertical sync or a monochromesignal detect is not falsely registered, the two most significantdigital video bits must be simultaneously high for two sample periodsbefore these detect signals are registered and used by the encoder toreset the vertical counter 86 or to disable the reception of color. Inother words, the signal detects indicate, to a high probability, thatthe respective vertical synchronization or monochrome information ispresent in the composite video signal, but the absence of a signaldetect does not necessarily indicate that the respective information wasabsent.

The time-warp and segment scrambling or encoding method performed by thevideo scrambler 27 is illustrated in FIG. 9. The video portion 100 of ahorizontal line is subdivided into eight sectors of approximately 64video samples per sector. Each sector 101 is in turn subdivided into twosegments 102 of approximately 32 video samples per segment.

The segments in each sector are scrambled in accordance with three bits(I, R, S) generated by the random number generator 28, 48. The first twobits (I, R) may change only once every 32 video samples or once for eachsegment, while the third bit (S) may change once every 64 video samples,or once for each sector. The first bit (I) designates whether thepolarity of the corresponding segment is inverted. The second bit (R)designates whether the time sequence of the corresponding segment isreversed or inverted. The third bit (S) indicates whether the twosegments in the corresponding sector are swapped. The eight combinationsobtained when the first two bits are zero for the first correspondingsegment SEGMENT O) are shown in FIG. 9. Note that when the randomnumbers are all zero, the encoded sector is the same as the originalunencoded sector. It is also evident from FIG. 9 that there arethirty-two different possible scrambled combinations for each sector,corresponding to the combination of the five different scrambling bits(First I&R, Second I&R, Sector S) indicated in FIG. 9. Note that in themethod as shown in FIG. 9, the samples in a given sector are scrambledonly with other samples in the sector.

The random number for the scrambling method of FIG. 9 is easilygenerated by a random number generator of the type shown in FIG. 10. Aset of four-bit shift registers 103 is loaded with the four-bit portionsof the seed number as the housekeeping line is received. The two bitsfrom each digital sample, obtained from the two most significant bits ofthe digital video, are multiplexed to four bits by a two bit latch 103aclocked at the sample clock frequency divided by four or approximately2.7 MHz. A random number bit is generated by key logic 104 from theoutputs of the four-bit shift registers. The key logic 104 alsogenerates an input to the shift registers 104 which is serially shiftedthrough the shift registers by one bit position at the boundary of eachsector. The key logic 104 preferably is a combinatorial logic network orPLA. A PLA having a reasonable number of gates generates a "pseudo"random number providing sufficient security.

As was the case with detection of vertical sync and the monochrome bit,it is important that the seed number should be latched into the shiftregister 103 only when it is certain that the seed number has beenreceived correctly. Otherwise the random number will be incorrect untilthe next frame when the seed is again loaded. To substantially eliminatethe falsing problem, each four-bit nibble of the seed is transmitted induplicate and each nibble is loaded into the shift register 103 onlywhen the duplicates match. An error detecting circuit 105 has a four-bitregister 105a for temporary storage and a digital comparator 105b fordetermining whether the previously received four bits match the currentfour bits fed to the shift register 103. AND gates 105c permit the shiftregisters 103 to be loaded only when the magnitude comparator 105bindicates a match.

According to another feature of the present invention, the undesirableeffects of discontinuities at the segment boundaries in the scrambledvideo signal are suppressed during encoding by inserting between thesegments, an additional "consecutive" sample at the beginning and end ofeach segment in the scrambled signal, the samples being consecutive inthe order of the samples in each segment in the scrambled signal. Beforedecoding, the inserted samples are deleted from the received encodedvideo.

Consider first the simplified method shown in FIG. 11. The transmittedencoded composite video 106 will have a discontinuity generallydesignated 107 at the boundaries between pairs of segments that are nolonger in their original time sequence after scrambling. Thus thereceived signal 108 will slew at the discontinuity 107 due to thelimited band width of the channel. At a sampling rate of 10.7 MHz and aband width of 4.2 MHz, the adjacent samples are not independent. Thus,the slew will generate an error in the sample 109 immediately followingthe discontinuity 107. One method to reduce this error is to insert asample at the beginning of each segment, the sample being set equal tothe first transmitted sample in each segment. Then, at the receiver, theinserted sample is disregarded because it is erroneous due to theslewing. It has been found, however, that this method is not entirelysuccessful due to the fact that the slope or time rate of change of thereceived and band-limited scrambled signal is not preserved at the endof the first segment and the beginning of the second segment in eachsegment pair.

The error generated by slewing at the discontinuities is virtuallyeliminated by a preferred method shown in FIGS. 11A-11C. The preferredmethod inserts an additional "consecutive" sample at the beginning andend of each segment in the scrambled signal, the respective samplesbeing consecutive in the order of the samples in each correspondingsegment in the scrambled signal, to thereby preserve the time rate ofchange of the received and band-limited scrambled signal at the end andbeginning of each segment in the band-limited scrambled signal.Referring now to FIG. 11A, an unscrambled video signal generallydesignated 111 is repetitively sampled, for example by the eight-bitanalog-to-digital converter 25 in FIG. 1, generating an ordered sequenceof original samples S₀ -S₁₇. Suppose, for the sake of argument, that thevideo signal 111 is subdivided into three segments 112, 113, 114, eachsegment containing six samples, for the purpose of scrambling by timeinversion of the second segment 113. In general, the signal is generatedby scrambling the segments of original samples so that the sequence oforiginal samples in the scrambled signal is not consecutive betweenadjacent segments at segment boundaries. According to the preferredmethod of the present invention, the effect of any discontinuity atsegment boundaries is reduced by inserting in the scrambled video signal111 in FIG. 11B two additional samples for each segment boundary, one ofthese samples being added to each adjacent end of the adjacent segmentsat the segment boundaries. The inserted samples are S₆ ', S₁₂ ', S₅ 'and S₁₁ ', having values equal to the respective values of the originalsamples S₆, S₁₂, S₅, and S₁₁ that are consecutive with the orederedsequence of original samples in the respective segments to which theadditional samples are added. In other words, the inserted samples areconsecutive in the order, either forward or reverse, of the respectivesegments to which they are added in the scrambled signal 111'. Thus atthe end of the first segment 112 there is inserted the additional sampleS₆ ' to be consecutive with the forward order of samples S₀ -S₅ in thefirst segment 112. Additional sample S₁₂ ' is inserted at the beginningand additional sample S₅ ' is inserted at the end of segment 113 to beconsecutive with the reverse order of samples S₁₁ -S₆. Additional sampleS₁₁ ' is inserted at the beginning of the third segment 114 so as to beconsecutive with samples S₁₂ -S₁₇.

Due to the effect of the inserted samples S₆ ', S₁₂ ', S₅ ' and S₁₁ ',the band-limited scrambled signal 111" in FIG. 11C has its time rate ofchange preserved at the beginning and end of each segment; in otherwords, at samples S₅, S₁₁, S₆, and S₁₂. Hence all of the originalsamples in the unscrambled video signal 111 are faithfully reproduced inthe band-limited scrambled signal 111". The inserted samples S₆ ', S₁₂', S₅ ', and S₁₁ ', however, are not reproduced in the band-limitedscrambled signal 111". They are, in effect, sacrificed to preserve thetime rate of change at the beginning and end of each segment.

An additional improvement is obtained by inserting a third sample ateach segment boundary. The third sample is inserted between the insertedsamples added to the ends of the adjacent segments and preferably has avalue set equal to the average of the inserted samples added to the endsof the adjacent segments. The average is calculated, for example, by abinary adder fed by latches gated to receive the values of the insertedsamples added to the ends of the adjacent segments. A right displacementof the adder outputs provides the required division of two. In FIG. 11B,the third samples are designated 115 and 116; a time scale expansion toaccommodate the third samples is presumed. The third samples 115, 116tend to compensate for any excessive deviation of the inserted samplesS₆ ', S₁₂ ', S₅ ', and S₁₁ ' away from the adjoining segments. The thirdsample 115, for example, tends to cancel the downward curvature of thescrambled signal 111' at the inserted sample S₁₂ '. The third sample 116tends to smoothly join the inserted sample S₁₁ ' and the inserted sampleS₅ '. In practice, the insertion of third samples is required only ifthe transmission channel is poor, for example, if ringing is excessive.

The technique of inserting samples during encoding and deleting thesamples during decoding adds complexity due to the change in the lengthof the video portion of the horizontal line. If the line length wereconstant, the segment swapping and time inversion could be performedwith two 64-byte memories. Then the sectors would only need to bealternately loaded into one or the other of the memories, and read outof the same memory with the memory addresses exclusive-OR'ed by the tworandom number bits (R, S). But the change in video line length requiresthe reading or writing of the video samples to be controlled inaccordance with the particular sector involved, as is illustrated inFIG. 12 for the worst case situation of segment swap and time inversion.

As is evident from FIG. 12, coded sequence 121 must be initiallydisplaced from the unencoded sequence 122 by a delay of 64 samples.Thereafter, the displacement increases by two samples for every32-sample segment. It is also evident that at least 128 bytes of memoryare required for the encoding process, since some samples are delayed byat least 128 sample times during encoding. Moreover, it appears thatmore than 128 bytes of memory are required to prevent a memorycontention problem during overlap intervals designated 123. If thestream of unencoded video samples were merely loaded into sequentialmemory addresses, then segment 4 would, at a later time, occupy the samememory addresses as segment 0, and segment 8 would, at a later time,occupy the same memory addresses as segment 4. But then the overlapintervals 123 would cause errors in the encoding process, since sampleswould be written over and destroyed before they were read out of thememory. The first five samples of segment 8, for example, would bewritten over the first five samples of segment 4 before the first fivesamples of segment 4 are read out of the memory. As is evident from thedecoded sequence 124, this memory contention problem also exists duringthe decoding process due to overlap intervals 125.

Shown in FIG. 13 is a block diagram of a video scrambler/descrambler 27,47 employing a 256 byte memory 130 in order to avoid the memorycontention problem. A latch 131 receives the digital video samples andfeeds the samples to the memory 130. After the time delays indicated inFIG. 12, the samples are read from the memory and received by a latch132. A bank of exclusive-OR gates 133 selectively complements the videosamples in order to perform the polarity inversion scrambling functionin response to the random number bit (I). The selectively complementedvideo samples are received by an output latch 134 having tristateoutputs which are enabled to perform an output multiplexing function (34in FIG. 1).

Since the segment swap, time invert, and polarity invert scramblingfunctions are complementary, the same components make up the videoscrambler 27 and descrambler 47. The only difference between thescrambler 27 and descrambler 47 is the position of a memory read-writeswitch or mask-programmed junction 135, the position T being selectedfor scrambling and the position R being selected for descrambling.During scrambling the digital video samples (122 in FIG. 12) aresequentially written into the memory 130, and later read from memorywith extra samples inserted (121 in FIG. 12). During descrambling thedigital video samples with extra samples inserted (121 in FIG. 12) arewritten into the memory and later sequentially read from the memorywithout extra samples (124 in FIG. 12). The change in position of theswitch 135 from scrambling to descrambling reflects the fact thatread/write function is complementary for scrambling versus descrambling.The read/write input to the memory 130 is either in phase with the 10.7MHz sample clock or 180° out of phase, as provided by an inverter 136.Thus the cycle time of the memory is on the order of 45 nanoseconds asdigital video samples are alternately written into and read from thememory 130.

The sequential addresses for the memory 130 are provided by a latch 137receiving a "dithered" horizontal count. The latch 137 addresses thememory 130 whenever unencoded video samples (122 in FIG. 12) are writteninto the memory or when decoded video samples (124 in FIG. 12) are readfrom the memory. Another latch 138 is provided to supply nonsequentialaddresses for reading or writing the scrambled video samples (121 inFIG. 12). The latches 137, 138 have tristate outputs that are wiredtogether to provide an address multiplexing function. The outputs of thelatches 137, 138 are alternately enabled at the 10.7 MHz frequency ofthe sample clock.

To generate the nonsequential addresses corresponding to the timepositions of the scrambled video samples, an eight-bit counter generatesa monotonic address sequence which is scrambled in accordance with therandom number bits (S) and (R). A bank of exclusive-OR gates 141selectively complements the five least significant outputs of thecounter 139 in order to perform the time inversion function, in responseto the random number bit (I). A single exclusive-OR gate 140 selectivelycomplements the counter output A₅ to perform the segment swap function,in response to the random number bit (S).

In order to duplicate the desired samples in successive segments duringthe encoding process, the eight-bit counter 139 is decremented at thesegment boundaries. For this purpose a decrement gate signal from avideo scrambler PLA 143 is applied to the decrement or "up/down" inputof the counter 139. The counter 139 is initially reset by a start videogate signal from the video scrambler PLA 143. During encoding the memory130 is read twice using repeated address after the counter 139 isdecremented, so that the inserted samples have the same values as thesamples that are consecutive with the first and last samples in the32-bit segments. During decoding the eight-bit counter 139 is inhibitedby a hiccup gate from the PLA 143 as the encoded sequence 121 is writteninto memory so that these pairs of inserted samples are written to thesame address locations in the memory 130. The inserted samples, in otherwords, are deleted by being written over in the memory.

The video scrambler PLA 143 receives the dithered horizontal count whichis generated by a video scrambler horizontal counter 144. The videoscrambler horizontal counter 144, like the horizontal counter 83 in thesystem timing circuit (FIG. 8) is preset by the horizontal reset signaland is clocked at the 10.7 MHz sample clock rate. The video scrambler'shorizontal counter 144, however, has a random initial state specified bya four-bit random number RND₃₋₀ generated by the pseudo-random numbergenerator 48 (FIG. 2). The positions or segment boundaries at which thevideo line is subdivided into segments are dependent on the initialstate of the video scrambler horizontal counter. In other words, thevideo lines are subdividied at different predetermined positions orhorizontal counts in the video lines for different video lines. Therandom initial state has the effect of randomly distributingdiscontinuties over the video line so that visual perception ofdistortion introduced by discontinuties is further suppressed.

The method of encoding the stereo audio onto the composite video signalis shown in FIG. 14. Each of the two audio channels are sampled anddigitized three times, and the audio bits for each channel aresequentially accumulated according to the schedule 160, for each 63.5microsecond interval of the horizontal line 165. The digital audio 166is inserted in 36 sample clock cycles between the chroma burst andvertical sync 168, and the scrambled video 169. Each sample 171 of thetwo channel digital audio is encoded as three analog samples 172, 173,174 per horizontal line, each analog sample having a time duration oftwo sample clock periods. The first two analog samples 172, 173 peraudio sample 171 are quantized to four levels, providing the four mostsignificant bits for the audio sample. These four most significant bitsare easily scrambled to scramble the audio sample 171. The third analogsample 174 or remainder is quantized to 256 levels, providing the eightleast significant bits of the 12 bits per audio sample 171.

The audio encoder is shown in FIG. 15. The two channel audio inputs21a', 21b' are fed to respective 12-bit analog-to-digital converters31a, 31b. The digital-to-analog converters 31a, 31b sample therespective audio inputs 21a', 21b' at the respective load times 160(FIG. 14) as determined by sequential and periodic load signals LA-LF.Respective OR-gates 175a, 175b clock or strobe the respectiveanalog-to-digital converter 31a, 31b upon the occurrence of load signalsLA, LC and LE, or LB, LD and LF, respectively. The least significanteight bits from the analog-to-digital converters 31a, 31b are fed torespective triples of eight-bit latches 176a, 176b. The four mostsignificant bits from the analog-to-digital converters 31a, 31b arereceived and scrambled by respective sets of four exclusive-OR gates177a, 177b in response to random number bits RND₃₋₀. The two mostsignificant bit outputs of the sets of exclusive-OR gates 177a, 177 bare fed to respective triples of 2-bit latches 178a, 178b and the twoleast significant bit outputs of the sets of the exclusive-OR gates177a, 177b are also fed to respective triples of two-bit latches 179a,179b. The latches 176a, 176b, 178a, 178b, 179a, 179b are clocked orstrobed at respective load times 160 by the indicated register loadsignals LA-LF and the latches present output signals to a tri-statedriver 180 at register transmit times 166 in response to the indicatedregister transmit signals TA-TR. The tri-state driver 180 transmits thescrambled digital audio over video bus lines V₇ -V₀ to the eight-bitdigital-to-analog converter 33 in FIG. 1.

It should be noted that the times for the sampling, loading, andregister transmitting occur periodically at the horizontal scanfrequency. Hence, the required times are obtained from the horizontalcount via a programmable logic array 181.

Generally speaking, each channel of the station audio 21a', 21b' issampled by a respective one of the 12-bit analog-to-digital converters31a, 31b, the four most significant bits of the 12-bit audio samples arescrambled by the exclusive-OR gates 177a, 177b, and the digital samplesare shifted in time from the register load schedule 160 to the registertransmit or digital audio schedule 166 by the registers 176a, 176b,178a, 178b, 179a, and 179b. Each analog sample having a time duration oftwo sample clock periods, corresponding to the respective registertransmit times 166 in FIG. 14, is temporarily stored for reformatting ina respective one of the registers in FIG. 15 being output enabled by therepective register transmit signal TA-TR.

It should be noted that the two-bit latches 178a, 178b hold the firstanalog sample 173, the two-bit latches 179a, 179b hold the second analogsample 173, and the eight-bit latches 176a, 176b hold the third analogsample 174, for each audio sample 174. It should be noted, however, thatthe quantized levels in the first two analog samples 172, 173 do notmatch or align with the quantized levels for the two most significantbits of the eight bits encoded in the third analog sample 174. An offsetis intentionally introduced so that it is easy to regenerate at thedecoder 40 the two bits corresponding to the four levels of the firsttwo analog samples 172, 173. The regeneration is accomplished bytruncation of the six least significant bits at the output of theeight-bit analog-to-digital converter 45 (FIG. 2). The offset isintroduced by a tristate driver 182 enabled whenever a first or secondanalog sample 172, 173 is transmitted as enabled by a NOR gate 183receiving the transmit signals TC, TF, TI, TL, TO, TR for the eight-bitregisters 176a, 176b. Thus the tristate driver 181 is enabled when thefirst two analog samples 172, 173 are transmitted and is disabledwhenever an eight-bit analog sample 174 is transmitted. A secondtristate driver 184 sets the lower five video bus lines V₄ -V₀ to logiclow when the two-bit analog samples 172, 173 are to be transmitted. Itshould be evident that the tristate drivers 182, 184 cannot be enabledsimultaneously with the outputs of the eight-bit registers 176a, 176bsince the outputs are wired together.

The audio descrambler 52 is shown in FIG. 16. It should be evident thatthe audio descrambler 52 uses essentially the same components as theaudio scrambler 32 (FIG. 15) and the wiring in FIG. 16 is drawn inmirror image relation to the wiring in FIG. 15. The main functionaldifference is that the registers in the audio descrambler 52 are clockedby the respective register transmit signals TA-TR and are output enabledby the respective load register signals LA-LF. In other words, the wiresto the clock input C and output enable input OE for each register arereversed to wire the registers for either an audio scrambler 32 or anaudio descrambler 52. The audio descrambler, however, uses twelve-bitdigital-to-analog converters 53a, 53b instead of the analog-to-digitalconverters 31a, 31b.

In general terms, the audio descrambler 52 receives the digital audio166 from the composite video signal 155 (FIG. 14) in digitized form fromthe digital video bus leading from the eight-bit analog-to-digitalconverter 45 in FIG. 2. The registers 176a, 176b, 178a, 178b, and 179a,179b are sequentially clocked or strobed by the register transmitsignals TA-TR (FIG. 14) so that the binary values of the analog samples172, 173, 174 of the audio samples 171 are latched into the respectiveregisters in response to the respective register transmit signals. Foreach register load signal LA-LF shown in FIG. 14, a particular set ofthree registers holding the three analog samples 172, 173, 174corresponding to a particular one of the six audio samples for thehorizontal line is output enabled by a respective one of the sixregister load signals LA-LF to generate twelve binary bits fed to arespective one of the two digital-to-analog converters 53a, 53b. Thefour most significant bits, however, are first descrambled by arespective one of two sets of four exclusive-OR gates 177a, 177b, inresponse to the random number bits RND₃₋₀. The respectivedigital-to-analog converter 53a, 53b receives the 12 bits and generatesan analog signal fed to either the channel one or the channel two audiooutput, when the respective digital-to-analog converter is activated bya respective clock signal. The clock signals are provided by NOR gates175a, and 175b activated by the respective combinations of load signalsLA, LC, LE, or LB, LD, LF. The register transmit signals TA-TR and theregister load signals LA-LF, are provided by the programmable logicarray 181 in response to the horizontal count.

A schematic diagram for the chroma-burst phase-locked loop and systemclock is shown in FIG. 17. A Schmitt trigger 211 is biased between themaxima and minima of the chroma burst in the composite video, therebygenerating a 50 percent duty cycle binary representation of the chromaburst. An exclusive-OR gate 212 compares the phase of the compositevideo to the phase of the 3.58 MHz output of a voltage controlledoscillator 210. An analog transmission gate 213 enabled by the chromaburst gate selects the phase error due to the chrome burst in thecomposite video. A demodulator low pass filter comprising a resistor 214and a capacitor 215 suppresses the seven MHz product signal, therebydetecting the phase error between the chroma burst and the 3.58 MHzoutput of the voltage control oscillator. A value of 22 K ohms for theresistor 213 and a value of 1000 picofarads for the capacitor 214 arerepresentative.

A noninverting operational amplifier 216 acts as a buffer and is alsobiased for a gain of approximately 20. A feedback resistor 217 of value100 K ohms and a resistor 218, of value 5.1 K ohms, provides therequired negative feedback. The output of the amplifier 216 is then fedto the voltage controlled oscillator 220 to complete the chroma burstphase-locked loop.

The voltage controlled oscillator 210 is crystal controlled at thesystem clock frequency of six times the chroma burst frequency. A divideby six counter 220 provides the 3.58 MHz VCO output fed to the phasecomparator 212. The crystal 221 has its resonant frequency pulled by avaractor 222 reverse biased by a resistor 223 typically of value 100Kohms. The varactor 222 is loosely coupled to the crystal 224 through acapacitor 224 typically of value 50 picofarads. The cathode of thevaractor 222 is shunted to signal ground at the 21 MHz frequency by acapacitor 225, typically of value 1000 picofarads. The gain for thecrystal oscillator is provided by a transistor 229 having an emitterload resistor 230 and biasing resistors 231 and 232. Typical componentvalues are 2.2 K ohms for resistor 230, 22 K ohms for resistor 231, and11 K ohms for resistor 232. A capacitor 226 typically 100 picofarads anda capacitor 227 typically 10 picofarads provides the positive feedbackfrom the emitter to the base of the transistor 229. The 21 MHz signal istapped off the emitter and fed to a buffer 234. A second buffer 235provides the 21 MHz system clock to the encoder or decoder. A two stagebinary counter 236 provides the 10 MHz sample clock and the five MHzclock, which are obtained by dividing the system clock by two and four,respectively.

The divide by six counter 220 is comprised of three J-K flip-flops 238,239 and 240 clocked by the 21 MHz clock. The K inputs of flip-flops 238and 239 are tied to logic high. The J input of flip-flop 238 receivesthe inverted Q output of flip-flop 239, and the J input of flip-flop 239receives the Q output of flip-flop 238 so that a division by threeoccurs. The third J-K flip-flop has its J and K inputs both tied to theQ output of flip-flop 239 to provide an additional division by two. Bydividing the 21 MHz frequency of the crystal 224 by six, the 3.58 MHzVCO output for the phase detector 212 is obtained. Since an exclusive-ORphase detector 212 is used, the divide by six counter 220 has beendesigned to give a symmetrical or 50% duty cycle output. A buffer 241provides a 3.58 MHz output for regenerating the chroma burst for thecomposite video transmitted from the encoder to the decoder, or for theNTSC video transmitted from the decoder to the customer's televisionset.

According to another feature of the present invention, a horizontal syncdetector 82 (FIG. 8) is provided which can accommodate various linelengths and can tolerate missing horizontal sync pulses. If, forexample, the horizontal sync detector 82 were merely a counter whichrolled over upon a predetermined count, a variable line length could notbe accommodated so long as the line length exceeded the roll over countof the counter. Moreover, in response to missing horizontal sync pulses,the counter would always roll over on a fixed count which would bedifferent than the count required to accommodate a variable line length.In other words, the horizontal counter 83 should be periodically reseteven though horizontal sync pulses are missing, and the frequency of thehorizontal counter reset must be variable, in response to the frequencyof the horizontal sync pulses when the horizontal sync pulses werereceived.

Shown in FIG. 18 is a circuit schematic for the horizontal sync detector82 which performs the above described functions. In simplified terms,the horizontal sync detector 82 determines whether the period betweenthe horizontal preset pulses is within a predetermined time window, andif so, the period is memorized so that the horizontal preset pulses maybe regenerated in the case of missing horizontal preset pulses. Shown inFIG. 19 are the wave forms for the horizontal preset pulses 71b and awindow gate signal 250 which sets upper and lower bounds on thepermissible horizontal period. Preferably the horizontal preset pulseshave a width of one system clock cycle, or approximately 50 nanoseconds.Increased precision is obtained by running the horizontal sync detector82 at the 20 MHz clock frequency instead of the 10 MHz sample clockfrequency.

The window gate 250 is generated by the combination of a synchronous upcounter 251 and a window PLA 252 which generates the window gate 250when the output of the up counter is within a predefined numericalrange. The up counter 251 is comprised of three eight bit synchronouscounters, part No. F196. The horizontal preset signal 71b presets the upcounter 251 to an initial state of value 104. The counter continuescounting until the output Q₁₁ goes high, thereby inhibiting furthercounting via the low-order clock enable signal CEP0. Configured in thisway, the up counter 251 is a resettable one-shot. The window PLA 252generates the window gate signal 250 whenever the output of the upcounter 251 is within the range of 1440-1480. An AND gate 253 combinesthe horizontal preset with the window gate to generate a qualifiedpreset. The qualified preset is less sensitive to noise since it occursonly when just two horizontal preset pulses have been received that areseparated by the horizontal frequency to within the window gate on time.The qualified preset is then used in the decoder in lieu of thehorizontal preset signal. The qualified preset signal could also be usedto gate the preset input to the counter 251. In such a case the counter251 must be permitted to freewheel so that the horizontal preset signalmay be at first acquired. Use of the qualified preset to gate the presetof the counter 251 could be desirable in certain "suppressed sync"systems wherein the horizontal preset 71b is obtained by detecting apseudorandom sequence which frequently appears in the digital data orvideo portion of the horizontal line. The desirability of feeding backthe qualified preset to gate the preset enable of the counter 251 shouldbe established by simple experimentation. It should also be noted thatthe window PLA 252 conveniently generates the chroma burst gate duringcounts of 130 to 184 of the up counter 251. Because the chroma burstphase-locked loop 43 can reject noise, it does not matter that thechroma burst gate is occasionally erroneous due to false triggering ofthe up counter 251.

The qualified preset is used to preset a down counter 254 whichregenerates the horizontal preset for missing qualified preset pulses.The down counter 254 is preset to a specific value depending on theremainder from the up counter 251 upon the occurrence of the qualifiedpreset so that the period of the regenerated horizontal preset pulses isprecisely the period of the horizontal preset pulses when the horizontalpreset pulses were present. In other words, the counters 251, 254 countin opposite directions so that the regenerated sync pulses are phaselocked to the input stream of horizontal pulses despite varying linelength. Preferably a gated latch 255 is used to remember the periodbetween the horizontal preset pulses so that the horizontal presetpulses may be regenerated by the down counter 254 even when multiplehorizontal preset pulses are missing. Part No. F173 may be used for thegated latch 255. If a gated latch is used, the qualified preset signalis preferably delayed by a flip-flop 256 so that the remainder value istransferred to the down counter 254 immediately after it is determined.It should be noted that if a gated latch 255 is not used, the delayflip-flop 256 need not be used.

The down counter 254 receives the remainder value on its eight leastsignificant parallel inputs P₀₋₇ and receives an offset value of 1280 onits higher order parallel inputs P₈₋₁₁ so that the up counter 251 andthe down counter 254 will count through the same number of cyclesbetween qualified horizontal preset pulses. The final count of the downcounter 254 is indicated by the higher order carry-out signal TC-3 whichis active when the outputs Q₈₋₁₁ are zero, or, in other words, at state255. The carry-out TC-3 is fed back to the parallel enable input of thecounter 254 by combining the carry-out TC-3 with the qualified preset inan OR-gate 257. The parallel input to the up counter 251 and the offsetP₈₋₁₁ to the down counter 254 may be experimentally determined bychoosing those offsets which for the case of a periodic horizontalpreset signal, generate the carry-out TC-3 precisely time coincidentwith the qualified preset to the OR gate 257. The output of the OR gate257 is also used as the regenerated horizontal preset fed to thehorizontal and vertical counters 83, 86, respectively (FIG. 8).

For decoding the digital program and customer data, it is especiallyimportant that the time positions of the digital data be preciselydetermined by a horizontal preset pulse. Thus, the absence of aqualified preset must inhibit the gated latches which receive thedigital customer data for each horizontal line. For digital customer andprogram data, the data are periodically supplied to the decoders so thatthe receipt of the data will merely be delayed if the data latches areinhibited even for as long as a few frames. It is also desirable toinhibit the digital-to-analog converters 53 from responding to thedecoded audio data in the absence of a qualified preset.

To generate a qualified horizontal line signal for gating the customerand program data latches, a JK flip-flop 258 is set whenever a qualifiedpreset signal occurs and is reset whenever a carry-out TC-3 of the downcounter 254 occurs without a coincident qualified preset. The set or Jinput of the flip-flop 258 is, therefore, merely the qualified presetsignal and the K, or reset, input is generated by an AND gate 259.

Even though the regenerated horizontal preset is used to reset aseparate horizontal counter 83 (FIG. 8), it may be desirable to use theoutputs of the down counter 254 to specify operations occurring orsynchronized to the end of the horizontal line, since the output of thedown counter 254 in terms of counts from the end of a line issubstantially independent of line length. One such operation occurringat the end of a horizontal line is the generation of the groundreference gate. As was described in conjunction with FIG. 5 and to beshown below, the ground reference gate could be generated from theleading edge 60a of the horizontal sync tip. But for the case of "20IEEE suppressed sync," to be described below, an end of line PLA 260 ispreferably used to generate the ground reference gate. The max referencegate, as well as the ground reference gate, should be gated by thequalified horizontal line signal. Shown in FIG. 21 is an AND gate 261provided for qualifying the ground reference gate.

Shown in FIG. 20 is a schematic circuit diagram for the horizontal synctip window detector 81 (FIG. 8). The composite video is fed to aninverting amplifier 265 which raises the conventional one voltpeak-to-peak signal level to approximately five volts peak-to-peak. Thefive volt signal is fed to a self-adjusting threshold detector 266 whichstrips off the peak of the horizontal sync tip. The peak of thehorizontal sync tip is synchronized to the 21 MHz clock by a delayflip-flop 267. In order to detect the leading and trailing edges of thepeak signal, the output of the delay flip-flop 267 is fed to a seconddelay flip-flop 268 and AND gates 269 and 270 each compare an output ofthe flip-flop 267 to an output of the flip-flop 268 to generate thehorizontal preset and horizontal reset signals, respectively. Thehorizontal preset signal is generated from the AND gate 269 combiningthe inverted output of the flip-flop 267 and the noninverted output ofthe flip-flop 268. The horizontal reset is generated by the AND gate 270combining the noninverted output of the flip-flop 267 and the invertedoutput of the flip-flop 268. The horizontal reset, for example, is usedto reset a counter 271 which is self-inhibited upon reaching its higheststate 15. The inhibit signal and also the ground reference gate aregenerated by a PLA 272. The ground reference gate, for example, isgenerated for states 5-11 of the counter 271. The counter 271 is, forexample, a synchronous counter part No. F169.

The self-adjusting threshold detector 266 is shown in detail in FIG. 21and is explained in conjunction with the timing diagram of FIG. 22. Theself-adjusting threshold detector 266 receives the five voltpeak-to-peak inverted composite video generally designated 275 in FIG.22. The inverted composite video 275 has its maxima at the horizontalsync tip 276 which is preceded by a video signal 277 and followed by themax reference pulse 278. In terms of a standard composite video signal,the maximum peak-to-peak voltage of the signal is divided into 140 equalparts called IEEE units. The level of maximum brightness is at 100 IEEEunits and the horizontal sync pulse is at minus 40 IEEE units. Thehorizontal sync pulse 276 is most difficult to detect when the videosignal 277 is encoding a dark blue picture. At this time the videoportion 277 extends from approximately 45 to minus 15 IEEE units. Theoptimum threshold voltage V_(t) ' for the self-adjusting thresholddetector 266 is at approximately minus 27 IEEE units.

In accordance with another aspect of the present invention, theself-adjusting threshold detector 266 uses a single operationalamplifier to function both as a peak detector and as a thresholddetector having a threshold set in relation to the peak amplitude of thecomposite video signal. As is known in the art, a peak detectorcomprises an operational amplifier having the signal received on itspositive input and having its negative input shunted to ground through aholding capacitor. The holding capacitor is charged through adirectional diode from the output of the operational amplifier. Theself-adjusting threshold detector 266 in FIG. 21 also has an operationalamplifier 280, a directional diode 281, and a holding capacitor 282. Thecapacitor 282 typically is of value 0.6 microfarads. The five voltpeak-to-peak inverted composite video is fed to the positive input ofthe operational amplifier 280 through a series resistor 283 typically ofvalue 20 K ohms. But the capacitor 282 is not directly connected to thenegative input of the operational amplifier 280; rather, it is seriesconnected through a resistor 284 typically of value 20 K ohms. Moreover,the output of the operational amplifier 280 is fed back to the negativeinput through a resistor 285, typically of value 200 K ohms. With thesespecified values of resistance and capacitance, the circuit in FIG. 21also generates a PEAK signal 286 shown in FIG. 22 which is the signalthat would be generated by a threshold detector having its thresholdV_(t) set at the desired level. The circuit of FIG. 21 also works wellusing a current mirror amplifier such as part No. LM359.

The standard composite video signal as shown in FIG. 22 is quitewasteful in transmitted power or modulation index since signal levelsfrom about minus 15 IEEE units to minus 40 IEEE units do not carry anyvideo information but are merely used to facilitate the detection of thehorizontal sync pulse 276. In accordance with another feature of thepresent invention, a "20 IEEE" suppressed horizontal sync pulse 276' isused having a maximum value reduced to at least minus 20 IEEE units asshown in FIG. 23. Using the detector circuit in FIG. 21, the peak signal286' will have a series of pulses 287 at the 3.58 MHz frequency of thesuppressed chroma carrier for the worst case of a pure dark blue colorpicture. In other words, the extent of modulation in the minus IEEEunits direction may be established by the worst case video modulationrather than the necessity for the sync pulse to be at a more negativeIEEE level. Preferably, the horizontal sync pulse amplitude is limitedto the maximum permitted swing of the video modulation. A discriminatorcircuit shown in FIG. 24 can effectively eliminate the periodic pulses287 so that the suppressed horizontal sync tip 276' may be detected. Thepeak output of the threshold detector 266 is fed to the D input of afour stage shift register 291. The shift register 291 is clocked at the21 MHz clock rate. A peak signal without the pulses 287 may be obtainedby the logical AND of at least two outputs of the shift register 291.The exemplary horizontal preset detector circuit 290 in FIG. 24logically combines all four outputs of the four stage shift register 291using an AND gate 292. A horizontal preset output is convenientlyobtained from a D flip-flop 293 which receives the output of the ANDgate 292 on its D input and is clocked at the 21 MHz rate. The requirednegative transition (294 in FIG. 23) is selected by an inverter 295which receives the PEAK signal and has its output fed to the AND gate292 for combination with the outputs Q₀₋₃ of the shift register 291.

One of the advantages of the present invention is that the conventionalaudio carrier of the television channel has been eliminated, therebyeliminating adjacent sound channel interference. This is one factorwhich helps ensure that the discontinuities introduced by the videoscrambling method will not be visible in the descrambled picture.

In accordance with another feature of the present invention, the digitalcustomer and program data is encoded using a correlative signallingtechnique so that the digital data is more efficiently transmittedwithin the television channel. Up to now it has been assumed that twobinary bits are transmitted for every two video samples at a 10.7 MHzvideo sample rate. In view of the Nyquist criteria, it should bepossible to transmit digital samples at the 5.4 MHz digital sample ratesince the television channel has a maximum band width of 4.3 MHz. Inpractice, however, the intersymbol interference between the digitalsamples is objectionable. Although the transmission of four-bit digitalsamples at a 2.7 MHz rate will eliminate the intersymbol interference,it is not the most efficient and practical method for transmitting thedigital data. It is preferable to live with the objectionableintersymbol interference at the 5.4 digital sample rate. A promisingalternative is to use a correlative signalling technique whereinintersymbol interference is deliberately introduced so that, in effect,the spectra of the digital signal more closely matches the frequencyresponse of the transmission channel. Correlative signalling techniquesare discussed at length by Dr. Adam Lender in "Correlative (PartialResponse) Techniques And Applications To Digital Radio Systems,"reprinted in, Feher, Digital Communications Microwave Applications,Prentice-Hall Inc., Inglewood Cliffs, New Jersey, 1981, pp. 144-182.

For the present video transmission system, the data are efficientlytransmitted at a 10.7 megabit per second rate using a duobinarycorrelative encoding method discussed in Example 7.1 on page 153 of theFeher text. A practical circuit is shown in FIG. 25. Assuming that thepreviously described digital data formatting circuits provide two bitsD₁ -D₀ at the 5 MHz clock rate, the correlative encoder 300 needs amultiplexer 301 for alternately selecting either the most significantbit D₁, or the least significant bit D₀ at the five MHz clock rate togenerate a single bit data stream at approximately 10 MHz. These singlebits at 10 MHz define two levels. In accordance with the correlativesignalling method, these two levels are differentially encoded using amodulo two adder and then the differential data is linearly superimposedor convolved over two 10 MHz sample intervals. An exclusive-OR gate 302performs the single-bit modulo two addition. The modulo two sum isdelayed in a delay flip-flop 303 clocked at the 10 MHz sample clockrate. The delayed modulo two sum is fed back as one input to the adder302 and added to the output of the multiplexer 301.

The modulo two sum from the adder 302 and the register 303 are summedtogether in a second adder generally designated 304 thereby generatingthe correlative encoded data which defines three levels. The threelevels are specified by hexadecimal values of FF, 3F and OO on theeight-bit video output V₇ -V₀. The most significant bit V₇ is a logicalone only when both the input D and the output Q of the delay flip-flop303 are high. Thus the most significant bit V₇ is provided by an ANDgate 305. The less significant bits V₆ -V₀ are all logical zero onlywhen both the input D and the output Q of the delay flip-flop 303 arelow. Thus, the less significant bits are provided by a NOR gate 306 andan inventer 307.

It should be noted that the correlative encoded values can be thought ofas the superposition of two data streams that change at a five MHz ratebut which are offset from each other. Each of these data streams,however, can pass through the video transmission channel as well as theunencoded data consisting of four bits at a five MHz rate. But thetransmitted correlative encoded data is quantized to three levelsinstead of the four levels required to quantize the unencoded data.Thus, it is expected that the data should be more reliably detected atthe decoder because the three levels are spaced further apart than thefour levels required for transmitting the unencoded data. Theoretically,the increase in transmission efficiency is obtained due to the fact thatthe spectra of the correlative encoded data is a better match to thefrequency response of the television channel. Unlike the spectra of theunencoded data which is relatively flat up to approximately 2.5 MHz anddrops off rapidly above 2.5 MHz, the spectra of the correlative codeddata gradually drops off between 2.5 MHz and 5 MHz. In both cases,however, the binary data is transmitted at the same density of about onebit every 100 nanoseconds.

In the encoder 300 of FIG. 25, the output of the second adder 304 islatched in a register 308 having tristate outputs which are wired-OR'edto input bus of the the video digital-to-analog converter 33 in FIG. 1.The outputs of the register 308 are enabled by a transmit digital gatewhen the digital signal should appear in the encoded composite video.

A corresponding correlative decoder 310 is shown in FIG. 26. Since thenumber of unencoded levels is two, the decoded levels are obtained bytaking mod two of the encoded levels ranging from level zero to leveltwo. Thus, the occurrence of level one gives the decoded bits directly.The three encoded levels are obtained as the logical AND 311,exclusive-OR 312, or logical NOR 313 of the two most significant bits V₇-V₆ on the digital video bus which were generated by the decoder'sanalog-to-digital converter 45 in FIG. 2. The decoded bits at the 10 MHzrate are alternately latched into registers 314, 315 at the 5 MHzdigital sample clock rate.

Another advantage of using correlative coding is that errors may bedetected by determining whether the value of the received encoded bitsV₇, V₆ change in value by more than a value of one from one 10 MHzsample clock cycle to the next. From the circuitry of the encoder 300 inFIG. 26, it is apparent that the encoded duobinary cannot change betweenlevels zero and two from one 10 MHz sample clock cycle to the next. Themaximum change occurs, for example, when the register 303 holds a valueof one and a new value of zero is strobed in, or when the register 303holds a value of zero and a new value of one is strobed in. To determinewhether a change between levels zero and two has occurred, register 317receives the three signals indicating levels zero and two. A logicnetwork 318 comprising three NAND gates 319, 320, 321 determine whethera direct change between levels zero and two has occurred and if so, anerror indication is generated at the output of gate 320. This errorindication could be used, for example, to suspend the latching of newcustomer and program data, to squelch the audio descrambler byinhibiting the loading of the D/A converters 53 to hold the present12-bit audio samples or it could be used as a diagnostic tool monitoringthe performance of the video transmission channel.

The decoder 40 shown in FIGS. 2 and 8, has been described as having ahorizontal sync detector 82 operating at a frequency of 21 MHz buthaving a horizontal counter 83 and other system timing circuits 44operating at 10.7 MHz. It is evident that if the horizontal counter 83were merely reset upon the occurrence of the horizontal reset pulsegenerated by the sync detector 82 operating at 21 MHz, there would be aloss of precision since the horizontal sync pulse could occur duringeither the first half or the second half of a 10.7 MHz sample clockcycle, but the horizontal counter 83 would respond in the same fashionto either the "early" or "late" horizontal sync pulse. But by using a"re-phase" circuit shown and described in FIGS. 27 and 28, it ispossible to synchronize the horizontal counter 83 to a sync pulse toobtain a timing resolution exceeding the period of the clockingfrequency of the counter. Instead of using a faster clock to obtainincreased synchronization precision, the clock signal to the counter isselectively stretched by one-half cycle so that the counter may, ineffect, count on either the leading or trailing edge of the unalteredclock. The timing resolution is increased from one period to 1/2 of theperiod of the clock. The clock is selectively stretched 1/2 period uponreceipt of the sync pulse depending on whether the sync pulse isreceived during the first half or the last half of the clock period.

Turning now to FIG. 27, a 2 X CLOCK signal 330 is shown corresponding tothe 21 MHz clock used, for example, in the synchronization circuit ofFIG. 18, 20, or 24. The unaltered clock 331 corresponds to the 10.7 MHzsample clock shown clocking the horizontal counter 83 in FIG. 8. A syncpulse synchronized to the 2 X CLOCK 330 could either occur during aninitial half period 332 or a final half period 334 of the unalteredclock 331. Pulse A 335, for example, occurs during the initial halfcycle 332 of the unaltered clock 331. In response to pulse A, there-phased clock A 336 is the same as the unaltered clock 331. But inresponse to a pulse B 336 occurring during the second half period 334 ofthe unaltered clock 331, the rephased clock B 337 has a portion 338which is stretched by 1/2 period of the unaltered clock so that there-phased clock is phase shifted 180° with respect to the unalteredclock 331. Assuming that the horizontal counter 83 (FIG. 8) is clockedon the rising edge of the re-phased clock, it is evident that dependingon whether pulse A 335 or pulse B 336 is used to reset the horizontalcounter, the actual counting will be selectively delayed by 1/2 of theunaltered clock. Hence, the horizontal resolution of the horizontalcounter 83 in response to a sync pulse, is doubled without requiring theclocking frequency to be doubled.

A representative re-phase circuit 340 is shown in FIG. 28. The re-phasecircuit 340 receives an asynchronous peak signal having a high-to-lowtransition to which a horizontal counter is to be synchronized. Thispeak signal is received on the D input of a delay flip-flop 341 clockedby the 2 X CLOCK, for example at 21 MHz. Its Q output is delayed by asecond delay flip-flop 342 also clocked by the 2 X CLOCK signal. A NANDgate 343 is active low upon receiving a high Q output from a secondflip-flop 342 and a high Q output of the first delay flip-flop 241 tothereby sense the high-to-low transition in the PEAK signal. The outputof the NAND gate 343 provides the PULSE signal active low, which is fedto a second NAND gate 344.

The output of the NAND gate 344 is fed to the D input of a third delayflip-flop 343 also clocked by the 2 X CLOCK. The Q output of the thirddelay flip-flop 343 is fed back to a second input of the NAND gate 344so that the rephased clock, at 10.7 MHz, appears at the Q output of thedelay flip-flop 343.

In functional terms, in the absence of an active low PULSE signal, theNAND gate 344 functions as an inverter so that the complement of thestored value in the delay flip-flop 343 is fed back to the D input ofthe flip-flop. Thus, the flip-flop 343 functions as a single stagebinary divider, to generate a 10.7 MHz clock from the 21 MHz clock. Butif the PULSE signal becomes active low, then the NAND gate 344 isinhibited and presents a logic high signal to the delay flip-flop 343 sothat the re-phased clock will assume a logic high state for the nextcycle of the 2 X CLOCK signal regardless of its initial state. Note thatthis is a sufficient condition to guarantee that the re-phased clock 336or 337 will be generated in response to the PULSE signals 335 or 336,since the desired re-phased clock 336 or 337 always assumes a logic highafter the occurrence of either PULSE A 335 or PULSE B 336.

As shown in FIG. 8, the ground reference for the eight-bitanalog-to-digital converter 45 is obtained by shunting the receivedencoding composite video to ground upon the occurrence of a groundreference gate. It has been found that such a technique requires ratherwide horizontal sync tips in order for the ground reference to beproperly established. A circuit shown in FIG. 29 is preferably used topermit a ground reference to be established by narrow horizontal synctips such as are shown in FIG. 5. This clamp circuit generallydesignated by 350 in FIG. 29 receives the video input on a load resistor351 of approximately 82 ohms. The video input is coupled by a 100microfarad capacitor 352 and a 0.1 microfarad capacitor 353. A 500 ohmvariable resistor 35a in series with the capacitors 352, 353 provides again adjustment. The video signal is then fed through a 750 ohm resistor354b to the negative input of a differential amplifier 355. Thedifferential amplifier 355 has a 2.2 picofarad capacitor 356 and a 7.5 Kohm resistor 357 providing negative feedback. The positive input of thedifferential amplifier 355 receives the minimum value of the videosignal so that in effect the clamped inverted video appearing at theoutput of the differential amplifier 355 is obtained as the differencebetween the video input and the minimum value of the video input. Theclamp circuit, in other words, does not shunt the video input to signalground, rather, the clamp circuit generates a reference for thedifferential amplifier 355 so that the clamped inverted video has itsmaximum value set to a predetermined reference level. The maximum valueof the clamped inverted video is obtained by a Schotky diode 358charging a storage capacitor 359 of value 0.1 microfarad. A dischargepath for the capacitor 359 is provided by a resistor 360 of value 1megohm. The most positive value of the clamped inverted video becomesstored on the capacitor 359, and this maximum value is buffered by anoperational amplifier 361 using a 100 K ohm series input resistor 362and a 22 megohm negative feedback resistor 363. The positive input ofthe operational amplifier 361 receives a reference voltage of +5 voltson its positive input so that the maximum of the inverted video isclamped to +5 volts. The output of the operational amplifier 361 is fedthrough a 9.1 K ohm resistor 364 back to the positive input of thedifferential amplifier 355. Note then that there is a negative feedbackloop from the differential amplifier 355 through the diode 358 andthrough the operational amplifier 361. Although the clamp circuit 350 inFIG. 29 uses a diode 358 so as to clamp the maximum of the invertedvideo to +5 volts, the diode 358 may be replaced by a transmission gatesuch as the transmission gate 87b in FIG. 8, to clamp any particularportion of the video signal to the +5 volt reference level.

From the foregoing, it can be seen that the subscriber cable televisionsystem of the present invention has a decoder employing digitalcircuitry which may be embodied in low-cost large-scale integratedcircuits. The system, however, has extremely high security and also anincreased capacity for transmitting program and customer data to thedecoder units. For ease of data handling, two channel audio, video, andhigh capacity program and customer data are multiplexed for transmissionon the same composite video signal. The decoder unit employs a systemtiming circuit which precisely synchronizes the sample times on thereceived composite video signal to the chroma burst, regardless ofwhether the video information is for a color or black-and-white program.An improved time-warp and segment scrambling method was disclosedrequiring reduced memory requirements and a memory architecture isdisclosed for

What is claimed is:
 1. A method of transmitting an audio signalcomprising the steps of:(a) repetitively digitizing the audio signal torepetitively generate a digital portion and a respective remainderportion, wherein said digitizing is performed by sampling the audiosignal at spaced instants in time and quantizing each audio sample to acertain number of spaced levels, the digital portion being the quantizedaudio sample, the remainder portion being approximately the differencebetween the audio signal and the digital portion at said spaced instantsin time, (b) transmitting the digital portions and their respectiveremainder portions as a series of analog samples, the analog samplesrepresenting the digital portions being quantized to more than twolevels, the analog samples representing the remainder portions havingsubstantially more resolution than the resolution of the analog samplesrepresenting the digital portions, each remainder portion beingtransmitted as a single analog sample, (c) receiving the series ofanalog samples, quantizing the analog samples representing digitalportions to generate quantized values, and adding the quantized valuesto values of the respective analog samples representing remainderportions to thereby generate a series of values representing thetransmitted audio signal.
 2. The method as claimed in claim 1, whereinthe digital portions are scrambled by pseudorandom numbers beforetransmission, and wherein the digital portions are descrambled by thesame pseudo-random numbers after transmission.
 3. A method oftransmitting an audio signal comprising the steps of:(a) repetitivelydigitizing the audio signal to repetitively generate a digital portionand a remainder portion, the remainder portion being approximately thedifference between the audio signal and the digital portion at spacedinstants in time, (b) transmitting the digital portions and theirrespective remainder portions as a series of analog samples, the analogsamples representing the digital portions being quantized to more thantwo levels, the analog samples representing the remainder portionshaving substantially more resolution than the resolution of the analogsamples representing the digital portions, each remainder portion beingtransmitted as a single analog sample, (c) receiving the series ofanalog samples, quantizing the analog samples representing digitalportions to generate quantized values, and adding the quantized valuesto values of the respective analog samples representing remainderportions to threreby generate a series of values representing thetransmitted audio signal, wherein the step of digitizing includesdigitizing the audio signal to 12 bit resolution at said spaced instantsin time, each digital portion including the four most significant bitsand each remainder portion including the eight least significant bits.4. The method as claimed in claim 3, wherein each digital portion istransmitted as two analog samples, each analog sample being quantized tofour levels, and wherein each remainder portion is transmitted as asingle analog sample.
 5. The method as claimed in claim 3, wherein thedigital portions are scrambled by pseudo-random numbers beforetransmission, and wherein the digital portions are descrambled by thesame pseudo-random numbers after transmission.
 6. A method oftransmitting an audio signal in a composite video signal comprising thesteps of:(a) repetitively digitizing the audio signal to repetititivelygenerate a digital portion and a respective remainder portion, whereinsaid digitizing is performed by sampling the audio signal at spacedinstants in time and quantizing each audio sample to at least sixteenspaced levels to generate the digital portion, the digital portion beingthe audio sample quantized to four bits of resolution, the respectiveremainder portion having a value that is approximately the differencebetween the audio sample and the respective digital portion at therespective spaced instants in time, (b) transmitting the digitalportions and their respective remainder portions as a series of analogsamples interlaced with a video signal in said composite video signal,the analog samples representing the remainder portions havingsubstantially more resolution than the resolution of the analog samplesrepresenting the digital portions, wherein each digital portion istransmitted as two analog samples each of which is quantized to fourlevels, and wherein each remainder portion is transmitted as a singleanalog sample, (c) receiving the series of analog samples, quantizingthe analog samples representing the digital portions to generatequantized values, and adding the quantized values to values of theanalog samples representing the respective remainder portions to therebygenerate a series of values representing the transmitted audio signal.7. The method as claimed in claim 6, wherein the digital portions arescrambled by pseudo-random numbers before transmission, and wherein thedigital portions are descrambled by the same pseudo-random numbers aftertransmission.